This invention relates to a switching power supply for supplying DC stabilized voltages to industrial and consumer equipment.
A conventional switching power supply has the arrangement shown in FIG. 22. In FIG. 22, there is shown a DC power supply 1 which is obtained by rectifying and smoothing an AC voltage or formed of a battery. This DC power supply is connected between input terminals 2, 2' so as to supply a positive voltage to the input terminal 2 and a negative voltage to the input terminal 2'. Shown at 3 is a transformer which has one end of a primary winding 3a connected to the input terminal 2, the other end thereof connected through a switching element 4 to the input terminal 2', one end of a secondary winding 3c connected to an output terminal 11', the other end thereof connected through a diode 7 to an output terminal 11, one end of a bias winding 3b connected to the input terminal 2', and the other end thereof connected to a synchronizing oscillation control circuit 13. The switching element 4 is turned on and off when supplied at its control terminal with an on-off signal from the synchronizing oscillation control circuit 13, thereby interrupting the input voltage to the primary winding 3a. The synchronizing oscillation control circuit 13 operates to change the on-period of the element 4 by the output signal from insulating transmission means 14 such as photocouplers and keeps the off-state until the voltage across the bias winding 3b is reversed in its polarity. The rectifying diode 7 has its anode connected to one end of the secondary winding 3c and its cathode connected to the output terminal 11. Shown at 9 is a smoothing capacitor, which is connected between the output terminals 11, 11'. The rectifying diode 7 rectifies the voltage induced in the secondary winding 3c connected between the output terminals 11, 11' and the smoothing capacitor 9 acts to smooth the rectified voltage and to supply it to the output terminals as the output voltage. Shown at 15 is an error amplifier which compares a reference voltage 16 and the output voltage between the output terminals 11, 11', amplifies the compared signal and supplies it to the insulating transmission means 14. The insulating transmission means 14 acts to insulate between the primary and secondary windings and to transmit the signal from the error amplifier 15 to the synchronizing oscillation control circuit 13. The operation of this conventional example will be described below.
The input voltage supplied from the DC power supply 1 which is connected between the input terminals 2, 2', when the switching element 4 is turned on by the on-signal from the synchronizing oscillation control circuit 13, is supplied across the primary winding 3a of the transformer 3 in the on-period to allow a primary current to flow in the primary winding, this causing magnetic flux in the transformer 3 so that energy is stored in the transformer 3. At this time, a voltage is induced in the second winding 3c of the transformer 3 so as to bias the rectifying diode 7 in the reverse direction. When the switching element 4 is turned off by the off-signal from the synchronizing oscillation control circuit 13, a flyback voltage is generated in the primary winding 3a and at the same time, another flyback voltage is generated in the secondary winding 3a so as to forward-bias the rectifying diode 7. Therefore, the energy stored in the transformer 3 is released in the form of a secondary current in the secondary winding 3c. This current is rectified by the smoothing capacitor 9 to produce the output voltage which appears between the output terminals 11, 11'. When the energy stored in the transformer 3 is all released, the flyback voltages in the primary and secondary windings 3 a, 3c disappear, and ringing is caused by the resonant voltage depending on the inductance and distributed capacitance of each winding. A similar voltage is also generated across the bias winding 3b of the transformer 3, and tends to change to the polarity opposite to that of the flyback voltage. The change of polarity is transmitted to the synchronizing oscillation control circuit 13, again turning on the switching element 4. Repetition of these operations provides a continuous output voltage from the output terminals 11, 11'.
Moreover, the operation in which the output voltage is controllably stabilized will be described in detail with reference to FIG. 23. Shown in FIG. 23 at (a) is a voltage waveform V.sub.DS across the switching element 4, at (b) is a primary current I.sub.D which flows in the primary winding 3a, at (c) is a drive pulse waveform V.sub.G from the synchronizing oscillation control circuit 13 and at (d) is a secondary current waveform I.sub.O which flows in the secondary winding 3c. In this figure, the solid lines indicate a so-called over-load time in which a large amount of output current I.sub.OUT flows out of the output terminals 11, 11', and the broken lines indicate a so-called light-load time in which a small amount of output current I.sub.OUT flows out of the terminals. In general, the output current I.sub.OUT is expressed by ##EQU1## the output voltage V.sub.OUT by ##EQU2## the switching frequency .function. by ##EQU3## where N.sub.S is the number of turns of the secondary winding 3c, N.sub.P is the number of turns of the primary winding 3a, L.sub.S is the inductance of the secondary winding 3c, V.sub.IN is the input voltage supplied from the DC power supply 1, TON is the on-period of the switching element 4, T.sub.OFF is the off-period of the switching element 4 and T is the oscillation period.
The output voltage V.sub.OUT is compared with the reference voltage 16 in the error amplifier 15 and the compared result is supplied through the insulation transmission means 14 to the synchronizing oscillation control circuit 13, thereby controlling the on-period of the switching element 4. Therefore, the output voltage is always controlled to be constant with the on-period changed even though the output current I.sub.OUT and the input voltage V.sub.IN are changed. FIG. 23 shows such situations. However, since the change of the on-period results in the change of the off-period, the oscillation frequency .function. is also changed as is obvious from the figure. Moreover, in order to prevent the on-period from being increased without limit by a over-current due to short-circuiting of the output terminals 11, 11' or the like, it is necessary that the synchronizing oscillation control circuit 13 has the function of limiting the maximum on-period or the primary current.
In the conventional switching power supply shown in FIG. 22, however, the voltage and current waveforms to be supplied when the switching element 4 is turned on and off are simultaneously changed while crossing each other due to the gradients depending on the response speed of the switching element 4, thus causing a large switching loss. In addition, if the switching loss is decreased by increasing the response speed of the switching element 4, the voltage and current waveforms become steeper, leading to an increase of the switching noise and the voltage and current spikes which are supplied to the switching element 4.
Recently, to solve such problems, various resonant-type switching power supplies utilizing LC resonance are proposed in which an inductance and capacitance are inserted on the switching circuit.
FIG. 24 shows an arrangement of a conventional resonant-type switching power supply. In FIG. 24, like elements corresponding to those in FIG. 22 are identified by the same reference numerals and will not be described. Referring to FIG. 24, there are shown an inductance 45 which is connected in series between the input terminal 2 and the primary winding 3a of the transformer 3, and a capacitor 41 which is connected in parallel with the switching element 4. These inductance 45 and the capacitance 41 constitute an LC resonant circuit. Shown at 5 is a diode which is connected across the switching element 4. This diode has its anode connected to the input terminal 2' and its cathode connected to one end of the primary winding 3a, so that when the energy stored in the inductance 45 is released back to the DC power supply 1, a current can be flow in the winding even under the off-state of the switching element 4. Shown at 42 is a synchronizing oscillation control circuit which generates an on-off control signal to the switching element 4 and detects the current in the diode 5 so that the on-period of the switching element 4 is changed and that the off-period is maintained until a current flows in the diode 5.
FIG. 25 shows operating waveforms at various points. In FIG. 25, at (a) is shown the voltage waveform VDS across the switching element 4, at (b) a current waveform I'.sub.DS which flows in the switching element 4 and in the diode 5, at (c) a drive pulse waveform V.sub.GI from the synchronizing oscillation control circuit 42, at (d) a current waveform I'.sub.c flowing in the capacitor 41, at (i) a secondary current waveform I.sub.O flowing in the secondary winding 3c, and (j) an induced voltage waveform V.sub.O across the secondary winding 3c. As will be understood from the operating waveforms, energy is stored in the transformer 3 and inductance 45 in the on-period of the switching element 4, while in the off-period, the energy stored in the transformer 3 is released through the secondary winding 3c to the output terminals 11, 11' and the energy stored in the inductance 45 is released to charge the capacitor 41. As a result, a sine-wave voltage vibrating at a resonant frequency, f.sub.C .apprxeq.1/2.pi..sqroot.LC determined by the inductance value L of the inductance 45 and the capacitance value c of the capacitor 41, is generated across the capacitor 41. Moreover, the vibrating sine wave voltage vibrates about the sum of the input voltage of the DC power supply 1 and the flyback voltage induced in the primary winding 3a. Thus, the values of the inductances 45 and capacitor 41 and the on-period are set so that the amplitude becomes much larger than the sum of the input voltage and the flyback voltage, causing the voltage across the capacitor 41 to be negative. This negative-voltage period causes the voltage across the capacitor 41 to be zero and a current is flowed in the inductance 45 through the diode 5. At this time, the switching element 4 is turned on by the synchronizing oscillation control circuit 42, but no discharge current flows from the capacitor 41 to the switching element 4, thus the so-called zero-cross switching being caused in which the switching element is in the on-state with zero voltage applied there across. This operation is repeated to supply the output voltage to the output terminals 11, 11'. In this resonant-type switching power supply, the voltage waveform (current waveform or both voltage and current waveforms depending on the system) applied when the switching element 4 is turned on and turned off gently changes with a sine-wave gradient irrespective of the response speed of the switching element 4. Thus, even if the current waveform is abruptly changed, the switching loss is small and since the voltage waveform is a sine wave, the switching noise is very little. In this resonant-type switching power supply, however, when the switching element 4 is turned on, the above zero-cross switching must be made, otherwise the capacitor 41 would be shorted through the switching element 4, thus breaking the switching element 4, or the switching loss would be suddenly increased or the switching noise would be increased. Therefore, it is very difficult to control the output voltage to be always constant against a wide range of input voltage and output current variations while the zero-cross switching is being secured. Also, other effective means have not yet been found so far. In other words, the minimum value of the on-period is limited for assuring the zero-cross switching and the off-period TOFF is also limited since the resonant frequency f.sub.C is maintained to be constant for assuring the zero-cross switching, or the off-period must be restricted in the range of ##EQU4## Thus, the control characteristic is not wide enough in the pulse-width control. Moreover, in the conventional system shown in FIG. 22 the voltage (or current)waveform applied to the switching element 4 is rectangular, but in the resonant-type it is a sine wave, so that the peak value of the applied waveform increases. This follows that the switching element 4 is required to have a large capacity and that a large resonant current is flowed through the inductance 45 capacitor 41 and primary winding 3C to the DC power supply 1.
Moreover, FIG. 26 shows another conventional switching power supply arranged to have a plurality of secondary windings in the transformer 3 and rectifying and smoothing means provided in each of the secondary windings, thereby generating a plurality of output voltages. In FIG. 26, like elements corresponding to those in FIG. 22, are identified by the same reference numerals and will not be described. In FIG. 26, there is shown a second secondary winding 3f wound on the transformer 3. This secondary winding has its one end connected to an output terminal 62' and its other end connected to an output terminal 62 through a rectifying diode 60. There is also shown a capacitor 61 which is connected between the output terminals 62, 62' and serves to smooth the induced voltage in the secondary winding 3f and to supply the output voltage.
The operations of the error amplifier 15, reference voltage 16, insulation transmission means 14, and synchronizing oscillation control circuit 13 by which the output voltage V.sub.OUT 1 between the output terminals 62-62' is controlled to be stabilized have been mentioned with reference to FIGS. 22 and 23 and thus will not be described again. The output voltage V.sub.OUT 2 between the output terminals 11, 11' which is not directly controlled, or an un-controlled output is a voltage proportional to the voltage output VOUT 1, or is expressed as ##EQU5## The stability of the output voltage V.sub.OUT 2 is not sufficiently high because the output current causes voltage drops chiefly across the impedance of the secondary winding, the operating impedance of the rectifying diode and the leakage inductance between the secondary windings 3c and 3f. In the above equation, N.sub.S1 is the number of turns of the secondary winding 3f and N.sub.S2 is the number of turns of the secondary winding 3c.
In this conventional arrangement of FIG. 22, in order to control the output voltage to be constant against the variations of the input voltage and output current, it is necessary to greatly change the oscillation frequency. However, since the upper limit of the oscillation frequency is determined by the operating frequency of the switching element 4, the response of the synchronizing oscillation control circuit 13 and of the control system and so on, the lower limit of the oscillation frequency must inevitably be lowered to widen the control range. Thus, the transformer 3 and the rectifying and smoothing circuits in the secondary winding, since they are designed for the lowest oscillation frequency, become large in their size and capacity, thus making the power supply be large-sized and expensive. In addition, since the control loop responds late to the transient variations of the output current, the output voltage has a transient variation. Particularly when the output voltage is transiently increased, the charge on the smoothing capacitor 9 is discharged through the output terminals 11, 11' in the form of the output current. Therefore, the discharge speed is slow when the output current is small, and hence it takes a long time for the output voltage to be stabilized, or the response of the output voltage to the transient change is very poor. Moreover, in order to protect the short circuit between the output terminals and the output current from over-current, it is necessary to provide a circuit for limiting the maximum value of the on-period. Also, for the insulation between the primary and secondary windings the insulation transmission means 14 such as a photocoupler is required to transmit a control signal. These requirements make the circuit arrangement complicated thereby increasing the cost. Moreover, in order to reduce the switching loss occurring when the switching element 4 is turned o and off thereby to make the high frequency operation possible, the response speed of the switching element 4 must be increased, thus increasing the switching noise. Consequently, the noise filter to be inserted in the input and output terminals for the purpose of preventing equipment from interfering with noise becomes large-sized. In addition, since the surge voltage and current to be applied to the switching element 4 are increased, a switching element of which capacity is excessively large is required. To solve these problems, if the rapid change of the turn-off waveform is suppressed, the turn-off loss and snubber-circuit loss increased, thus lowering the efficiency to an impractical extent. Another method for reducing both the switching loss and the switching noise at a time, or such resonant-type switching power supply as shown in FIG. 24 has been proposed, but has a difficulty in the stabilizing control of the output voltage to achieve the zero-cross switching. Also, since the voltage waveform or current waveform to be applied to the switching element 4 is sine wave which increases the peak value, the switching element 4 must have a large capacity and thus cannot be improved much in efficiency due to a large resonant current.
Moreover, the switching noise is caused not only in the primary winding of the transformer, but also in the secondary winding. The switching noise in the secondary winding is large which is caused by the generation of ringing waveform due to the distributing capacitance between the leakage inductance (between the secondary winding and the primary winding a viewed from the secondary winding side) and the secondary winding, and by the recovery current of the rectifying diode, chiefly when the switching element 4 is turned off. To cope with these problems, a snubber circuit is connected across the secondary winding or the rectifying diode as in the primary winding. However, the snubber-circuit loss is increased and the current flowing in the snubber circuit is transmitted from the secondary winding of the transformer to the primary winding thus increasing the spike current to cause a large loss when the switching element 4 is turned off.
In the conventional arrangement having a plurality of secondary windings shown in FIG. 26, the detected and controlled outputs of the multiple outputs are always controlled to be stabilized, but the other un-controlled outputs are much affected by the variations of the output currents so as to slightly change. Particularly at the time of light load when the output current of the un-controlled output becomes small, inadequate coupling of the secondary winding of the transformer which contributes to the controlled output, to the other secondary windings which are associated with the un-controlled outputs will cause the un-controlled output to be easily affected by the spike voltage produced from the leakage inductance of the transformer so that the output voltage of the uncontrolled output is greatly increased. In this case, the greater the output current of the controlled output and the less the output current of the un-controlled output, the more the output voltage of the un-controlled output will increase. In order to prevent the output voltage from increasing at the time of light load on the un-controlled output, a resistor or the like is connected between the output terminals of the un-controlled output so that the bleeder current flows therein not to cause light load, or a regulator circuit is connected before the output. However, the bleeder current and the regulator circuit cause losses, thus lowering the efficiency of the switching power supply.